A new, compact, and dual-band dual-polarized duplex (D3) phased array architecture is proposed in this study. In contrast to studies reported previously, this design integrates four independent beamforming systems within a single printed circuit board (PCB), enabling the proposed 1 × 4 phased array to transmit or receive simultaneously vertically and horizontally polarized signals at 28 and 38 GHz, thereby supporting concurrent, dual-band, and dual-polarized four-beam operations. In addition, the exceptional frequency selectivity of the phased array facilitates frequency-division duplex operations. By adopting a brick-type architecture, the proposed phased array achieves two-dimensional scalability, which allows it to serve either as a standalone, small-scale phased array, or as a sub-block for larger-scale arrays. A novel, dual-polarized end-fire magnetoelectric dipole antenna was developed as the radiating element for the phased array. This antenna exhibits an impedance bandwidth of return loss below −10 dB across the frequency range of 24.8-40.3 GHz (47.6%), which represents one of the broadest operating bands reported for PCB-based, co-apertured, and dual-polarized end-fire antennas. Experimental validation of the fabricated phased array demonstrated that the two orthogonal polarizations could achieve beam-scanning ranges exceeding 90° and 60° at 28 and 38 GHz, respectively. The measured effective isotropic radiated power values exhibited distinct frequency selectivities between the two bands. To the best of our knowledge, this is the first demonstration of a D3 phased array that presents a promising solution for beyond fifth-generation (B5G) and sixth-generation (6G) millimeter-wave multi-standard systems.
The technological trajectory indicates that wireless networks beyond fifth-generation (B5G) and sixth-generation (6G) are capable for comprehensive commercial deployment in the near future, heralding an era of networking marked by unprecedented speed, extensive coverage, and enhanced intelligence. The application of B5G/6G technologies is anticipated to cover diverse and complex scenarios including vehicle-to-everything (V2X), smart venue/home, smart manufacturing, remote sensing, and smart city, supporting high-bandwidth and low-latency communications essential for applications such as telemedicine, virtual reality, and autonomous driving [[1], [2], [3], [4], [5], [6]].
Millimeter-wave (mmWave) communication is a pivotal element in the evolution of B5G/6G technology, capitalizing on the distinctive attributes of high-data transmission rates and broad bandwidths [[7], [8], [9]]. High-path loss and weak penetration capability during signal propagation are challenges that must be overcome to enhance the coverage and system reliability of mmWave communications [10,11]. Beamforming techniques have been demonstrated as effective countermeasures. Among these, active phased array technology has emerged as the predominant technology because it can electronically control the phase and amplitude distribution across array elements, thus facilitating rapid beam scanning and flexible adjustment, further improving the directionality and selectivity of communication systems [[12], [13], [14], [15], [16]]. Compared with passive beamforming methods, such as Butler matrices [17] and multibeam lenses [18], active phased arrays offer superior scanning agility, greater flexibility, and enhanced reliability.
Although conventional, single-band, and single-polarized phased arrays have been extensively investigated and commercially deployed in 5G applications, they fail to satisfy the increasing performance and cost–effectiveness requirements of the forthcoming B5G/6G. Under these circumstances, dual-band phased array technology has emerged and is regarded as a critical solution for future B5G/6G mmWave multi-standard systems [[19], [20], [21], [22]]. Existing dual-band phased-array implementations typically employ spatially separated radiating structures for distinct frequency bands [23], which reduce the aperture efficiency. Conversely, a single-aperture, dual-band phased array can operate in two or more frequency bands within a unified radiating aperture, thereby reducing physical dimensions and deployment costs. Current dual-band phased array implementations primarily follow two strategies: One involves the utilization of a duplexer to combine two channels of different frequency bands, whereas the other employs a wideband channel that can cover both bands. The latter prevents the simultaneous generation of independent beams for discrete bands, making them incompatible with frequency-division duplexing (FDD) requirements. Numerous research studies have been conducted on mmWave, dual-band, or wideband passive antenna arrays [[24], [25], [26]], and the studies concerning the realization of active dual-band phased arrays have gained increasing scientific attention recently.
Active phased arrays capable of covering multiple mmWave communication bands have been reported in previous studies [[27], [28], [29], [30]], and are predominantly realized based on broadband radiating elements and beamformers. Unfortunately, these phased arrays are limited to single-polarization operations. In contrast, dual-polarized communication systems have the potential to enhance the spectrum utilization efficiency and communication stability through polarity diversity. However, to the best of our knowledge, only one study [19] reported a dual-band dual-polarized phased array design. Despite the notable progress manifested by the findings of this study [19], it did not support FDD operations and independent beams in dual-band cases because dual-band signals shared the same port in the proposed design. A dual-band dual-polarized duplex (D3) phased array featuring the same radiating aperture but independent ports for each operating band and polarization can simultaneously achieve FDD operation and polarity diversity, thereby supporting concurrent, dual-band, and dual-polarized four-beam operations. As depicted in Fig. 1, these attributes of the D3 phased array endow it with the competence for multi-standard communication scenarios with multiple user equipment in B5G/6G, such as V2X, smart homes, and smart manufacturing. Notably, this architecture achieves substantial reductions in size and cost along with improved spectral efficiency, thus establishing an unprecedented solution in the field.
Despite considerable progress in dual-band and broadband dual-polarized antenna investigations, the systematic integration of these components in a multichannel beamforming system and maintenance of independent beam operation remains unresolved system-level challenges. The proposed D3 phased array must be designed as a modular unit accounting for stringent size constraints to ensure scalability for large-scale array deployment. The primary technical obstacle resides in the compact arrangement of up to four beamforming systems and numerous passive components within confined spatial dimensions, while preserving stable radiofrequency (RF) interconnections between the front-end circuits and radiating elements. Owing to their high level of integration and lightweight volume, tile-type architecture-based phased arrays are currently prioritized [[31], [32], [33]]. However, the implementation of tile-type multiband phased arrays is hindered considerably by spatial constraints and thermal-management challenges. To alleviate the active circuit density and relax printed circuit board (PCB) design constraints, brick-type architectures have gained widespread adoption [[34], [35], [36]]. In these architectures, passive components and active circuits can be orthogonally arranged relative to the radiating aperture on different dimensions, thereby enabling large-scale D3 phased array implementations.
To resolve these technical barriers, a compact D3 scalable phased array is proposed in this study for B5G/6G applications, demonstrating the first monolithic PCB integration of four independent beamforming systems, supporting concurrent, dual-band, and dual-polarized four-beam operation with a hybrid FDD/time-division duplexing (TDD) capability. By adopting a brick-type architecture combined with a symmetrical lamination design, the challenges of multiband isolation and high-density RF front-end layouts are effectively addressed. Furthermore, a novel, broadband, and dual-polarized end-fire magnetoelectric (ME) dipole antenna is proposed as the radiating element of the phased array, achieving an overlapped bandwidth exceeding 45% within a single aperture, providing new insights for mmWave, end-fire, and dual-polarized broadband antenna design. The dimensionally scalable construction of the D3 phased array enables its operation as either a standalone compact system or as a modular subblock for large-scale array configurations. To the best of our knowledge, this is the first reported D3 phased array implementation offering an innovative solution for B5G/6G mmWave multi-standard communications.
2. Architecture of D3 phased array
Fig. 2(a) depicts dual-band dual-polarized phased arrays employing spatially separated radiating apertures for distinct frequency bands, which result in an increased physical size and manufacturing cost, thereby posing difficulties in the deployment of many space-constrained applications. Fig. 2(b) shows a single-aperture, wideband dual-polarized phased array. Although it enables dual-band functionality within a unified aperture, this implementation requires broadband beamforming systems or chips [37,38]. More importantly, independent beam scanning at different bands cannot be accomplished, and only TDD operation between the two frequency bands is supported, constraining the spectral efficiency. In contrast, Fig. 2(c) shows the proposed single-aperture D3 architecture, in which a broadband antenna is adopted to transmit and receive signals in both bands, followed by a duplexer connected to the broadband antenna to split the signals into two independent frequency channels that are subsequently routed to the corresponding beamforming systems. This architecture allows the phased array to receive or transmit simultaneously either vertically polarized (V-pol) or horizontally polarized (H-pol) signals in both bands, supporting concurrent, dual-band, and dual-polarized four-beam operation and obtaining FDD functionality. It is noteworthy that compared with switches, the duplexers incorporated in the D3 phased array architecture achieve concurrent dual-band operation through frequency-domain isolation rather than time-domain switching, thereby supporting FDD without incurring throughput degradation associated with temporal control. Enhanced spectrum utilization and improved communication flexibility can be achieved using the proposed architecture, thus providing a novel solution for future B5G/6G multi-standard wireless communication systems.
The schematic of the D3 phased array implemented in this study is shown in Fig. 3. The phased array adopted a brick-type architecture and was implemented using a multilayer PCB process. Four dual-polarized end-fire antennas were seamlessly integrated along the lateral edge of the PCB to form a 1 × 4 linear array along the y-axis. The front-end circuits were meticulously laid out and routed along the z-axis. The beamforming systems and passive components for orthogonal polarization were partitioned across two PCB surfaces and confined within a 1 × 4 array grid for a compact design. The proposed phased array offers scalability in both the x- and y-axis dimensions, allowing for easy expansion to larger arrays such as 8 × 8, 16 × 16, or beyond. A detailed discussion of the scalability of the proposed phased array is presented in Section 4.
3. Phased array implementation
3.1. Wideband dual-polarized end-fire antenna
Various publications have reported on broadband dual-polarized boresight antennas [[39], [40], [41]], typically realizing overlapped bandwidths exceeding 40% with excellent in-band impedance matching and radiation performance. However, reports on dual-polarized end-fire antennas with an overlapping bandwidth above 40% are limited. To satisfy the design requirements of a brick-type D3 phased array, a novel, dual-polarized end-fire ME-dipole antenna covering multiple mmWave communication bands was developed.
Unlike traditional implementations of dual-polarized end-fire antennas that consider separate radiators [[42], [43], [44]], the proposed antenna utilizes a single radiator for dual-polarized end-fire radiation, thereby enhancing the integration density. The configuration of the proposed antenna element, which employs a multilayer PCB process and features three-dimensional ME-dipole structures integrated along the lateral edge of the PCB, is illustrated in Fig. 4(a). The antenna is assembled using two identical radiator boards and an intermediate feeding board. Metallized via arrays were adopted in Refs. [45,46] to establish vertical electric (E) dipoles and ground planes, achieving V-pol end-fire ME-dipole antennas with broadband characteristics. However, these metallic walls impede the formation of a continuous horizontal current on the radiator, making it challenging to realize H-pol radiation. Therefore, partial hollowing out and metallization of the PCB were adopted in the proposed antenna to guarantee the continuity of currents along both the vertical and horizontal directions. Specifically, two single-layer RO4350B substrates were processed by hollowing and metallization to form radiators, which were then assembled with a feeding board. The hollowed-out and metallized area of the PCB, as shown in Fig. 4(a), is presented as an L-shaped slot with a certain height, constituting the arms of the E-dipole for both polarizations, and the quarter-wavelength shorted patch of the magnetic (M) dipole for horizontal polarization.
Dual-polarized excitation is implemented by using orthogonally oriented Γ-shaped probes, with feeding networks optimized for cross-polarization isolation. The laminated structure of the feeding board is shown in Fig. 4(b) and the dimensions of the structures in each key metal layer are shown in Fig. 4(c). The eight-layer feeding board employs substrate-integrated coaxial lines (SICLs) spanning multiple metal layers, exhibiting symmetrical lamination to facilitate phased-array integration. The feeding probe and SICLs for horizontal polarization are distributed on layer M4, through which the H-pol radiation can be effectively excited, and symmetrical radiation patterns are ensured. The SICLs for vertical polarization were located on layer M7, and were connected to a metallized via that constitutes a vertical Γ-shaped feeding probe in conjunction with a square patch on layer M1. In addition, the metalized via of the vertical feeding probe was positioned at a distance from the feeding probe of the horizontal polarization on layer M4 along the z-axis direction, ensuring a satisfactory level of cross-polarization isolation. Four square patches positioned on layers M1 and M8 were combined with radiator boards to form quarter-wavelength shorted patches, completing the V-pol M-dipole configuration.
The simulated scattering (S)-parameters of the proposed dual-polarized end-fire radiating element are shown in Fig. 4(d), where broadband impedance matching for both polarizations and a high level of cross-polarization isolation are observed. Experimental validation of the proposed antenna element should be conducted to confirm its performance in subsequent active phased array configurations, which require structural modifications in the hollowed-out and metalized areas of the radiator boards to meet manufacturing constraints. As shown in Fig. 5(a), the modified structure incorporates π-shaped configurations formed by dual L-shaped metallized slots and vertical ground planes, while maintaining structural integrity through metallized via arrays in other areas. These modifications had minimal impact on the impedance matching and radiation performance of the antenna. Furthermore, to facilitate measurements, the feeding board was dimensionally adjusted with integrated SICL-to-mini-coaxial transitions, as shown in Figs. 5(b) and (c), providing vertical interconnects between the multilayer SICLs and vertical launches. Simulation results indicated that the optimized transition structures exhibited good impedance matching and low-insertion loss within a broad operating band.
Detailed photographs of the fabricated antenna element with mechanically using screws are shown in Fig. 5(d). The simulated and measured S-parameters are plotted in Fig. 5(e), with a measured overlapped bandwidth of 47.6% (24.8-40.3 GHz) for |S11| < −10 dB, fully covering the mmWave communication bands n257 and n260. In addition, the cross-polarization isolation is typically more than 20 dB within the operating band. Fig. 5(f) shows the plots of gain versus frequency, exhibiting horizontal and vertical polarization peaks at 5.94 and 6.77 dBi, respectively. Fig. 6 shows the normalized radiation patterns in the E-planes and magnetic planes (H-planes) that demonstrate stable end-fire characteristics. Under current technological constraints, achieving an overlapped bandwidth exceeding 45% presents a major technical challenge in mmWave dual-polarized end-fire antenna designs. Therefore, the characteristics of the developed ME-dipole antenna demonstrate the antenna’s strong suitability for the proposed D3 phased array, exhibiting an exceptional balance among the operation bandwidth, gain performance, integration level, and polarization characteristics. Nevertheless, the inherent architectural innovation of the proposed D3 phased array provides theoretical compatibility with alternative antenna configurations depending on the customized implementation requirements.
3.2. System design
Fig. 7 (a) shows a block diagram of the proposed D3 phased array. Four broadband dual-polarized end-fire ME-dipoles are linearly arranged to form a 1 × 4 radiating array with a center-to-center spacing of 5.5 mm (0.51λ0 at 28 GHz and 0.7λ0 at 38 GHz, where λ0 is the wavelength in free space). Broadband power dividers together with integrated 28/38 GHz bandpass filters were utilized to form duplexers interfaced by four independent beamforming systems dedicated to specific frequency-polarization combinations.
Beamforming systems utilized two types of commercial multichannel beamforming chips, MSTR102E and MSTR104E [47], both of which were fabricated using the SiGe bipolar complementary metal-oxide semiconductor (BiCMOS) process and assembled through wafer-level chip-scale packaging (WLCSP). The MSTR102E operated across the range of 24.25-29.50 GHz, integrating quad transceiver channels with 6-bit phase shifters and 5-bit attenuators per channel. In addition, the integrated RF switches in each channel facilitate the TDD operation for the phased array. The chip can be programmed using an external microcontroller via a high-speed serial-peripheral interface (SPI) to support precise beam scanning. The MSTR104E mirrors this functionality for operations in the range of 37.0-42.5 GHz. The detailed performance specifications of both chips are listed in Table 1. The D3 phased array was constructed using two MSTR102E chips and two MSTR104E chips.
The phased array integrates beamforming systems onto a multilayer feeding board with two symmetrical radiator boards, maintaining an identical construction as the individual antenna elements. Fig. 7(b) shows the laminated structure of the feeding board employing additional blind vias for simultaneous RF/digital/direct current (DC) interconnection. The settings of the dielectric substrates and prepreg layers were consistent with those of the antenna element shown in Fig. 4(b); therefore, these were not labeled. Beamforming systems that serve as orthogonal polarizations are deployed on opposite PCB surfaces, leveraging the lamination symmetry to streamline the layout and routing design. The SICLs connected to the H-pol probe on layer M4 transitioned to a grounded coplanar waveguide (GCPW) on layer M1 through a quasi-coaxial vertical transition structure [48,49]. Similarly, the SICLs in layer M7 transitioned to the GCPW in layer M8 and were then connected to the duplex circuits.
The 28 and 38 GHz chains of horizontal polarization between the duplex circuits and beamformers are both routed on layer M1 using GCPW, necessitating cross-layer routing to resolve spatial conflicts. As shown in Fig. 7(c), two quasi-coaxial transitions between layers M1 and M4 are introduced, the 28 GHz transmission lines are transitioned to layer M4 and routed for a distance before transitioning back to layer M1, while the 38 GHz chains are maintained on layer M1 in the form of GCPW. Because the metal layers M2 and M3 act as ground planes, and given the arrangement of ground-shorted vias, exceptional shielding performance can be achieved between transmission networks operating at different frequencies. As shown in Figs. 8(a) and (b), the optimized quasi-coaxial vertical transitions exhibited minimal insertion and return losses across a broad operating band. Vertical polarization follows analogous routing principles on the layer M8, with 28 GHz signals transitioning to SICLs on layer M7. This design can ensure isolation between the two polarizations and avoid an increase in the blind-via types; however, it leads to localized crossover structures requiring electromagnetic optimization. The modeling and simulation of this structure are illustrated in Fig. 8(c). The results indicate that the isolation between the two transmission networks is approximately 20 dB within the operating frequency band. The common ports of the four multichannel beamforming chips are connected to the GCPWs on layers M1 and M8 and then terminated by four surface-mounted subminiature push-on micro (SMPM) RF connectors.
The duplex circuit was composed of a single-stage broadband Wilkinson power divider and two commercially available surface-mounted filters made by low-temperature co-fired ceramic (LTCC) technology with different passbands. The configuration and simulated performance of the power divider are presented in Fig. 8(d), exhibiting an insertion loss in the range of 0.3-0.7 dB, and isolation responses better than 14 dB within the operating band.
4. Experimental verification and analysis
To validate the effectiveness of the proposed design, a prototype of the D3 phased array (Fig. 9) was fabricated and evaluated. Beamforming chips, SMPM connectors, LTCC filters, and components of supporting circuits, such as resistors and capacitors, were installed on both surfaces of the PCB using surface-mounted technology. The DC power supply and digital control connections were made using flexible printed circuit plugs, which were also surface-mounted on the PCB. Similar to the assembly process of the antenna element, two small radiator boards were fixed to the lateral side of the active feeding board with screws, achieving a compact size of 64.0 mm × 26.5 mm.
4.1. Beam-scanning performance
Discrepancies among different RF channels of phased arrays are unavoidable; these can introduce considerable inconsistencies in the phase and amplitude. Thus, calibration is essential before the measurement of the beam-scanning performance [50,51]. As shown in Fig. 10(a), the fabricated phased array was calibrated and evaluated using a Keysight N5225B PNA network analyzer and a broadband standard gain horn antenna in an anechoic chamber. The phased array undergoes a far-field over-the-air (OTA) per-channel calibration methodology to estimate and compensate for channel inconsistency [52], which employs a standard gain horn antenna as the calibration probe and requires only channel activation/deactivation control to acquire N measurement sets for the M-element array calibration (where N = M). The detailed calibration procedure is as follows:
(1) Channel activation and signal acquisition. The target channel is exclusively activated via the SPI, while the others are suppressed. The vector network analyzer transmits continuous-wave signals to the activated channel, and the amplitude and phase of the signals received by the horn antenna are recorded.
(2) Phase/amplitude state traversal. All the states of the 6-bit phase shifter and 5-bit attenuator incorporated in the target channel are traversed, and error maps are generated by comparing the measured and theoretical outputs.
(3) Error compensation. Pre-compensation for the phase and amplitude is implemented in the beamforming algorithms based on error maps. For instance, a +15° phase offset is applied when the measured 30° output of the phase shifter deviated from its theoretical value of 45°.
(4) Full array iterations. This procedure is repeated for all 16 channels to generate a global compensation table.
After calibration, a reduction in the phase error’s root-mean-square (RMS) value (from 106.3° to 5.4°) was demonstrated, and an improvement in the amplitude error’s RMS value (from 1.85 to 0.38 dB) was obtained, validating the effectiveness of the calibration methodology. The measurement setup of the far-field anechoic chamber is shown in Fig. 10(b). Considering the reciprocity of the antenna array, only the beam-scanning performance in the transmitting mode was measured.
The normalized beam-scanning simulation and measurement patterns are plotted in Figs. 10(c)-(f), where the simulated results are indicated by the dashed lines and the measured results by the solid lines. Seven beam states with scanning angles of 0°, 15°, 30°, and 45° were measured at 28 GHz for both polarizations. Meanwhile, five beam states with scanning angles of 0°, ±15°, and ±30° were verified at 38 GHz. The ±45° beam states at 38 GHz were omitted owing to performance degradation caused by the appearance of grating lobes. In the horizontal polarization case, the measured gain reductions relative to broadside beams were 1.7/2.1 dB when the beam-scanning angle was ±45° at 28 GHz, and were 2.5/2.0 dB when the beam-scanning angle was ±30° at 38 GHz. In the vertical polarization case, the gain decreased by 2.4/2.7 dB when the beam-scanning angle was ±45° at 28 GHz, and decreased by 2.6/2.2 dB when the beam-scanning angle was ±30° at 38 GHz. Despite minor asymmetries and distortions in the measured patterns, which are mainly attributed to non-ideal manufacturing factors and errors introduced during calibration, satisfactory agreement between the simulated and measured results confirmed the successful implementation of the proposed D3 phased array system and demonstrated four independently steerable beams.
4.2. OTA performance
The effective isotropic radiated power (EIRP) values at 1 dB compression point (P1dB) for the proposed phased array under different beam states were measured and calculated using a comparative method. A schematic explaining the relevant measurements is shown in Fig. 11(a). With the remaining transceiver chains held constant, an active phased array and a standard-gain horn antenna were used as the transmitting ends. A spectrum analyzer was employed at the receiving end to collect the power received by the horn antenna. The received power is denoted as Pra (or Prh) when the active phased array (or horn antenna) is used as the transmitting end, respectively.
where Pin represents the input power of the signal source set at P1dB of the phased array, and GAPA represents the transmitting link gain of the phased array. Lct and Lcr are the insertion losses of the cables at the transmitting and receiving ends, respectively; Ght and Ghr are the gain values of the transmitting and receiving standard gain horns, respectively. LOTA represents the spatial path loss, which can be verified using the Friis transmission formula [53].
${{L}_{\text{OTA}}}\left( \text{dB} \right)=32.4+20\log R+20\log f $
where R is the transmission distance and f is the operating frequency.
Based on the comparative method, the EIRP can be calculated as
The measured frequency responses of the EIRP are plotted in Figs. 11(b) and (c). Calibration of the broadside beams was performed for the four beamforming systems at different frequency points, ensuring an optimal EIRP in the boresight direction of the phased array during different beam states. Within the 26.5-29.5 GHz band, the EIRP peaks of 30.6 dBm (H-pol) and 31.4 dBm (V-pol) are achieved, exhibiting spectral fluctuations below 1.5 dB. Corresponding measurements in the 37-40 GHz band yielded maximum EIRP values of 27.6 dBm (H-pol) and 27.5 dBm (V-pol), with fluctuation amplitudes limited to 2.4 dB. Furthermore, the H-pol 28 GHz beam achieved an out-of-band attenuation greater than 37 dB in the 37-40 GHz band, while the V-pol 28 GHz beam achieved an out-of-band suppression larger than 28 dB in the same band. Two differently polarized 38 GHz beams can achieve an out-of-band suppression above 32 dB in the 26.5-29.5 GHz band. These measurements reveal consistent EIRP frequency trends associated with orthogonal polarizations in both operational bands. Stable EIRP values can be achieved in both bands while maintaining a high level of out-of-band suppression, which provides a solid foundation for the FDD operation of the phased array.
System wireless transmission experiments were conducted using a 64-quadrature amplitude modulation (QAM) signal with a bandwidth of 80 MHz. A photograph of the measurement setup is shown in Fig. 12(a). The proposed phased array operated in the transmitting mode, and a standard-gain horn antenna served as the receiver. The modulated signal was generated by a signal generator that activated the proposed active phased array. The signal received by the horn was acquired by a spectrum analyzer and uploaded to a personal computer for demodulation and equalization processing, after which the constellation diagram and error vector magnitude values were extracted. Fig. 12(b) illustrates the captured constellation diagram for the 64-QAM signal, verifying reliable wireless transmission performance across multiple beams with distinct polarizations and frequencies.
4.3. Independence of concurrent multibeam operation: Experimental verification and discussion
Conventional passive multibeam antennas (e.g., Butler matrices) and active multibeam systems (e.g., full digital beamforming arrays) typically generate multiple beams with identical polarizations and frequency bands. Consequently, when multiple beams are simultaneously activated, interbeam interference inevitably occurs, thus deteriorating the radiation performance manifested by (among others) pointing accuracy, beamwidth, and gain characteristics. In contrast, the four beams of the proposed D3 phased array operated at distinct frequency bands with orthogonal polarization. Benefiting from the high-polarization isolation and superior frequency selectivity designed for the system architecture, the four beams were theoretically characterized by exceptional independence. As evidenced by the EIRP results in Fig. 11, interband beam suppression exceeding 32 dB was achieved, confirming excellent spectral isolation that ensures the independent operation of frequency-division beams. However, the polarization-domain independence between the co-band beams requires additional experimental validation.
Under full-polarization excitation conditions, the beam-scanning characteristics of the H-pol 28/38 GHz beams were experimentally verified using the setup illustrated in Fig. 13(a). The H-pol ports of both the phased array and the standard-gain horn antenna were connected to a vector network analyzer, whereas the V-pol ports of the phased array were driven by a signal generator to activate the V-pol beamforming. The steering performance of H-pol beams was characterized using a vector network analyzer. Reciprocal experimental configurations were implemented for the V-pol beams, as depicted in Fig. 13(b), with procedural details omitted owing to measurement symmetry. The normalized radiation patterns for all four beams under full-polarization excitation conditions are shown in Figs. 13(c)-(f). The results indicate that the H/V-pol beams can independently scan at distinct angles when dual-polarization links are concurrently activated, exhibiting performance consistency with single-polarization operations.
The frequency responses of the EIRP under full-polarization excitation were investigated further, as shown in Fig. 14(a). Both the H/V-pol ports of the phased array were simultaneously fed by signal generators, with all beamforming systems activated to generate broadside beams. During the measurement of horizontal/vertical polarization, the H/V-pol port of a standard-gain horn antenna was connected to a spectrum analyzer, thereby measuring corresponding polarization-specific EIRP responses. As shown in Figs. 14(b) and (c), the measured EIRP results demonstrate consistent power stability within the operational bands and sustained high-interband suppression, confirming negligible cross-polarization interference.
These findings collectively verify that the four beams of the D3 phased array can realize simultaneous beam generation and independent scanning capabilities, thus enabling high-operational independence.
4.4. Performance comparison and discussion
Table 2 [19,20,27,28,31,32] summarizes the performance comparison outcomes of the proposed D3 phased array and several recently reported, state-of-the-art, mmWave phased arrays. Although a 1 × 4 array paradigm is used for comparisons, the proposed phased array integrated multiple independently adjustable beams within a single array aperture, a capability not possessed by the large-scale phased arrays reported in Refs. [31,32]. In addition, the proposed phased array is scalable in both dimensions, enabling deployment flexibility across diverse application scenarios. Unlike the broadband phased arrays discussed in Refs. [27,28], the proposed architecture implemented frequency-selective beamforming through discrete mmWave band separation, permitting independent beam adjustments per frequency chain. In addition, the absence of dual-polarization capability in previous studies [27,28] was also realized. Most importantly, compared with the work reported in Refs. [19,20], the proposed phased array possesses dual-band and dual-polarization functions and incorporates four independent, concurrent, adjustable beams, and FDD capability. In addition, the operation mode of the TDD can be supported by built-in transceiver channels and RF switches. Compared with the more compact tile-type architecture, this design utilizes a brick-type architecture to accommodate four independent beamforming systems on a single board, while retaining the advantages of modular design, ease of maintenance, and better heat dissipation. Overall, the proposed phased array is the first demonstration of a single array aperture wherein dual-band, dual-polarization, and duplex functions are integrated, featuring compact size, stable gain, concurrent four-beam operation, and satisfactory frequency selectivity, making it a promising candidate for B5G/6G mmWave multi-standard communication systems.
4.5. Scalability demonstration
The 1 × 4 D3 phased array proposed in this study demonstrates excellent beam-scanning capabilities, rendering it suitable for short-distance communication scenarios. Whereas mmWave coverage expansion requires array dimensional scaling, the presented architecture achieved two-dimensional scalability without requiring modifications to existing active circuitry or passive component designs. Specifically, along the y-axis, the array can be expanded by increasing the PCB size to accommodate a larger antenna array and a supplementary active beamforming system. Scalability along the x-axis was achieved by arranging multiple identical PCBs in parallel and mounting them onto a unified rear support structure, thereby enabling a planar array extension.
The scalability was verified based on electromagnetic simulations of an upscaled 8 × 8 planar configuration, as shown in Fig. 15. The baseline 1 × 4 array was first extended along the y-axis, followed by a parallel arrangement of eight 1 × 8 subarrays along the x-axis with a 5.5 mm vertical spacing. The simulated beam-scanning performances of the two orthogonal polarizations in the xoz- and yoz-plane at 28/38 GHz are illustrated in Fig. 16. At 28 GHz, seven beam states that possess scanning angles of 0°, ±15°, ±30°, and ±45°, are displayed, while at 38 GHz, five fixed beam states that possess scanning angles of 0°, ±15°, and ±30°, are presented. It can be observed that the beam scanning ranges can fully cover the angular range of −45 to 45° at 28 GHz and −30 to 30° at 38 GHz with scanning losses below 3 dB for the two polarizations in both the xoz- and yoz-plane. These results confirm the preserved radiation characteristics when scaled to large planar configurations, validating the two-dimensional scalability of the proposed D3 phased array.
5. Conclusions
A D3 phased array was proposed and implemented for B5G/6G mmWave, multi-standard communication systems. Four independent beamforming systems were integrated on a unified board, enabling multibeam operation across distinct frequency bands and polarizations, while also supporting TDD/FDD functionalities. By properly arranging the beamforming systems and RF circuits utilizing multilayer PCB processing, compact size and scalability in two dimensions were realized. A prototype was fabricated and measured, and the measured results confirmed excellent beam-scanning performance at 28 and 38 GHz, regardless of horizontal or vertical polarization. In addition, excellent EIRP and out-of-band suppression performances were achieved among different frequency bands. Furthermore, the novel, dual-polarized end-fire ME dipole antenna reported in this study achieved a broadband overlapping bandwidth of exceeding 45% by using an innovative radiator structure design covering multiple mmWave frequency bands. Broadband impedance matching and high cross-polarization isolation characteristics facilitated the realization of high-performance D3 phased array. Compared with other mmWave phased arrays in the literature, the proposed method stands out for its distinct advantages in terms of multifunctionality, integration, and scalability.
The authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.
Acknowledgments
This work was supported in part by the National Science Foundation of China (62301152 and 62188102), the Natural Science Foundation of Jiangsu Province (BK20230819), the Fundamental Research Funds for the Central Universities (2242022k60003), and the Youth Talent Promotion Foundation of Jiangsu Science and Technology Association (TJ-2023-074).
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